NPN Common Emitter Bias
in [Design]
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From: Winfield Hill on 6 Jan 2006 09:01 James Arthur wrote... > > Winfield Hill wrote: > <snip> > >> The circuit has three problems, all easily fixed. [...] > > Here's a version that makes sure the inductor is bypassed at > RF frequencies, yet, per Win, avoids AM modulation of the bias > from power supply noise: > >. .---------------------+----------+------- +Vcc >. | C1 | | >. R1 100n _|_ Rs >. 300mV --- 200mV >. | R3 | R4 | >. v\| .--/\/\--+--/\/\----+--------. >. |---, | 50mV 50mV | | >. /| | |/v | --- C2 >. +-----+----| C| --- 1n >. | Q2 |\ C| L1 | >. R2. | C| === >. 5mA .-. | GND >. etc | | R5 | >. | | | 1k +------------ out >. GND '-' | >. | Q1 |/ >. RF IN------+-----------------| >. |\v >. | >. === >. GND This circuit has two problems, easily fixed. First, the 100mV drop across R3 and R4 is determined by the base current of Q1, the exact value of which is unknown, and which can change by a factor of say 3 over temperature, etc. So if the drop varies from say 50 to 150mV, then the current-sense voltage will vary from 250 to 150mV, which means we haven't done a very good job of setting Q1's collector current. To solve this we need to stabilize Q2's current. We can do this with a Q1 base-to-ground resistor, R6 = Vbe/I, sized to draw say 3x more current than Q1's base. We can hope this new resistor, R6, will have a higher value than Q1's RF input impedance. The second issue is loop stability. The loop gain is roughly gm1*Rs, which is 40*0.2=8, times R6/(R3+R4) = 750mV/100mV=7.5, for a DC gain of about 60. Capacitor C1 must provide a dominant pole, reducing the loop gain to below unity before the occurrence of a second pole. The second pole could be due to Rs C2, or due to the input coupling capacitor, C3, with R6. Either way, C1 will probably need to be larger, likely an electrolytic. Here's the new circuit: .. .---------------------+----------+------- +Vcc .. | C1 | | .. R1 elec _|_+ Rs .. 300mV --- 200mV .. | R3 | R4 | .. v\| .--/\/\--+--/\/\----+--------. .. |---, | 50mV 50mV | | .. /| | |/v | --- C2 .. +-----+----| C| --- 1n .. | Q2 |\ C| L1 | .. R2. | C| === .. 5mA .-. | GND .. etc | | R5 | .. | | | not too +------------ out .. GND '-' big | .. | Q1 |/ .. RF IN ---||---+----+------------| .. C3 | |\v .. R6 | .. | === .. GND GND I question the need for R5, which at any rate must not be too big, or drop more than a few volts. I question it because Q2's collector is a high-Z current-source output, most likely with a low capacitance, much smaller than Q1's base. But, if it's not too large, it won't hurt anything. :-) It could be an RFC. -- Thanks, - Win
From: dagmargoodboat on 6 Jan 2006 15:19 Winfield Hill wrote: > James Arthur wrote... > > > > Winfield Hill wrote: > > <snip> > > > >> The circuit has three problems, all easily fixed. [...] > > > > Here's a version that makes sure the inductor is bypassed at > > RF frequencies, yet, per Win, avoids AM modulation of the bias > > from power supply noise: > > > >. .---------------------+----------+------- +Vcc > >. | C1 | | > >. R1 100n _|_ Rs > >. 300mV --- 200mV > >. | R3 | R4 | > >. v\| .--/\/\--+--/\/\----+--------. > >. |---, | 50mV 50mV | | > >. /| | |/v | --- C2 > >. +-----+----| C| --- 1n > >. | Q2 |\ C| L1 | > >. R2. | C| === > >. 5mA .-. | GND > >. etc | | R5 | > >. | | | 1k +------------ out > >. GND '-' | > >. | Q1 |/ > >. RF IN------+-----------------| > >. |\v > >. | > >. === > >. GND > > This circuit has two problems, easily fixed. > > First, the 100mV drop across R3 and R4 is determined by the base > current of Q1, the exact value of which is unknown, and which can > change by a factor of say 3 over temperature, etc. So if the drop > varies from say 50 to 150mV, then the current-sense voltage will > vary from 250 to 150mV, which means we haven't done a very good > job of setting Q1's collector current. To solve this we need to > stabilize Q2's current. We can do this with a Q1 base-to-ground > resistor, R6 = Vbe/I, sized to draw say 3x more current than Q1's > base. We can hope this new resistor, R6, will have a higher value > than Q1's RF input impedance. > > The second issue is loop stability. The loop gain is roughly > gm1*Rs, which is 40*0.2=8, times R6/(R3+R4) = 750mV/100mV=7.5, > for a DC gain of about 60. Capacitor C1 must provide a dominant > pole, reducing the loop gain to below unity before the occurrence > of a second pole. The second pole could be due to Rs C2, or due > to the input coupling capacitor, C3, with R6. Either way, C1 > will probably need to be larger, likely an electrolytic. > > Here's the new circuit: > > . .---------------------+----------+------- +Vcc > . | C1 | | > . R1 elec _|_+ Rs > . 300mV --- 200mV > . | R3 | R4 | > . v\| .--/\/\--+--/\/\----+--------. > . |---, | 50mV 50mV | | > . /| | |/v | --- C2 > . +-----+----| C| --- 1n > . | Q2 |\ C| L1 | > . R2. | C| === > . 5mA .-. | GND > . etc | | R5 | > . | | | not too +------------ out > . GND '-' big | > . | Q1 |/ > . RF IN ---||---+----+------------| > . C3 | |\v > . R6 | > . | === > . GND GND > > I question the need for R5, which at any rate must not be too > big, or drop more than a few volts. I question it because Q2's > collector is a high-Z current-source output, most likely with a > low capacitance, much smaller than Q1's base. But, if it's not > too large, it won't hurt anything. :-) It could be an RFC. > > > -- > Thanks, > - Win Nice tweaks Win. My actual circuit, as I recall now some 16 years later (there were a few versions), was as Andrew's, with a 1-2V drop across the sense resistor, minimizing the temperature drift problem. A Miller integrating capacitor across the current-sense transistor ensured stability. The goal was to overcome the output transistor's native 6:1 beta range (at 25C); confining Q1's idle current to a 2:1 range was considered excellent control...the customary, competing circuit was--get this--a fixed resistor to Vcc! Even more surprising to us non-rf-types, was that 6:1 idle current variation wasn't really all that bad--the actual variation was somewhat less in practice, it worked fine, and only changed the r.f. output by say +/- 30%, i.e. just a dB or so. I wanted to save battery power, which was at a premium, and avoid modulating the transistor reactances and thus interfering with my i/o network optimizations. Of course the details of desired power output and supply voltage (dictating choice of Q1), input and output loading, and so forth are vital to any particular implementation. Here's the rationale for R5: In my case Q2 was a 2n3906, which shows about 2pF Ccb _typical_ with a few volts Vce in the Motorola Small Signal Transistor book. That 2pF was about double the _max_ input capacitance of Q1 alone, and added about 90 ohms' reactance in parallel with Q1's base--enough to detune its carefully tuned input. Isolating Q1(b) from Q2's collector with an inductor was tempting, except that inductors of the d.c.-choke scale were really just capacitors in sheep's clothing at 900MHz, having unspeakable parasitics which loaded the junction and added both resonances, and magnetic-coupling from L1 (bad!!) to the mix. Although simply strapping Q2(c) to Q1(b) and redesigning the input network might've been okay, it was easier, cheaper (several million of these devices were ultimately built) and sleep-at-night-happier to insert an R5 dropping a volt or two--no more--and thus be sure of being isolated, yet reactance and coupling-free. SMD resistors are surprisingly good at UHF. Cheers, James Arthur
From: Winfield Hill on 6 Jan 2006 15:35 dagmargoodboat(a)yahoo.com wrote... > > Winfield Hill wrote: >> I question the need for R5, which at any rate must not be too >> big, or drop more than a few volts. I question it because Q2's >> collector is a high-Z current-source output, most likely with a >> low capacitance, much smaller than Q1's base. > > Here's the rationale for R5: > In my case Q2 was a 2n3906, which shows about 2pF Ccb _typical_ > with a few volts Vce in the Motorola Small Signal Transistor book. > That 2pF was about double the _max_ input capacitance of Q1 alone... Whoa, interesting, what part was Q1? -- Thanks, - Win
From: dgc on 6 Jan 2006 22:07 "Ian Bell" <ruffrecords(a)yahoo.com> wrote in message news:dpk53e$1j3$2(a)slavica.ukpost.com... > John Larkin wrote: > >> On Thu, 05 Jan 2006 21:13:45 +0000, Ian Bell <ruffrecords(a)yahoo.com> >> wrote: >> >>>Jim Thompson wrote: >>> >>>> >>>> AND: TC of output DC bias point = -2mV/?C * DCGAIN >>>> >>> >>>Quite right, and probably the reason it is rarely used. In simple >>>applications where temperature range is limited it may not be a problem. >>>With a 12V supply and a 6V collector voltage the temp co is about >>>20mV/degree which over a 20 degree range is just 400mV. >>> >>>Ian >> >> Well, just run the low side of the bottom resistor to a negative >> voltage. That's a great way to bias gaasfets, too. >> >> John > > I suspect if the OP had a negative voltage available he would want the > output to swing all the way down to it ;-) > > Ian Ian I am the OP of this post and, after reading all the replies, am certain I am corresponding with people whose knowledge beta is well beyond mine. Appoligies for leaving out critical data in my post. The circuit is indeed RF (7 MHz). I am trying to squeeze 2 watts rms out of a two stage arrangement the first of which is a crystall oscillator and the final an NTE235 NPN in common emitter arrangement. I am trying to maintain linearity in the output waveform and my old BKPrecision oscilloscope indicates I am failing miserably at this. The drive from the oscillator is running about 1 volt peak (i know this is high) which may be part of the problem. The drive is brought in from a 1 turn winding off the oscillator tuned circuit torroid inductor. I had 500 mA of standing DC on the 235 at one point and did indeed achieve the best output waveform at this level of bias. The heat sinked 235 was still running pretty hot at this level so I backed off. I now have a larger heat sink (3 X 6) 1/32 aluminum which will be better I'm sure. Frankly I may have harmonics in the output waveform as well. I'm no expert in deciphering oscope waves unless they are pretty clean. Output load is 50 ohms, which if the equation V0^2 / 2Po is correct would indicate 50 ohms is too high for 2 watts (another problem). Would this be compounding the linearity issue? I got a kick out of your post. You bet, I'm the kinda guy that wants all the peak to peak voltage available from the supply.
From: dagmargoodboat on 7 Jan 2006 01:04
Winfield Hill wrote: > dagmargoodboat(a)yahoo.com wrote... > > > > Winfield Hill wrote: > >> I question the need for R5, which at any rate must not be too > >> big, or drop more than a few volts. I question it because Q2's > >> collector is a high-Z current-source output, most likely with a > >> low capacitance, much smaller than Q1's base. > > > > Here's the rationale for R5: > > In my case Q2 was a 2n3906, which shows about 2pF Ccb _typical_ > > with a few volts Vce in the Motorola Small Signal Transistor book. > > That 2pF was about double the _max_ input capacitance of Q1 alone... > > Whoa, interesting, what part was Q1? Q1 was an MRF571. Sweet part. You're right though, I mis-read the data sheet. C.in is actually about 1.4pF _typical_, which would initially seem to reduce the effect of adding 2pF by connecting Q2(c). That 1.4pF C.in, however, is reported at 1MHz and 5mA, and is not that experienced in actual operation. Under bias and 920MHz it looks to my addled rf-pate like the MRF571's base reactance is less capacitive, shifting to just about neutral or a little inductive, depending. Without recalling the exact particulars, the input matching network was a fixed network with values on the order of 2.2pF, 8.2pF, 5.6pF, etc., and had to be on-tune, so an extra 2pF was in any case unwelcome. And that, my friends, is the tale of the resistor R5, and how she came to be. Cheers, James Arthur (posted this earlier in the day, but it failed to show) |